The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.
Recently, due to the development of switching devices capable of withstanding relatively high currents and voltages of high frequency, most power supply circuits that rectify a commercial power supply voltage and obtain a desired direct-current voltage have become switching type power supply circuits.
Switching power supply circuits have a transformer and other devices miniaturized by raising the switching frequency, and are used as a power supply for various electronic devices, such as a high-power DC-to-DC converter.
Generally, when a commercial power supply voltage is rectified, a current flowing in a smoothing circuit has a distorted waveform, thus causing a problem in that a power factor indicating the efficiency of use of the power supply is degraded. In addition, a measure to suppress the harmonics caused by such a distorted current waveform is required.
So-called wide range-ready power supply circuits are known as switching power supply circuits configured to be able to perform operations dealing with a range of alternating input voltages of about AC 85 V to 288 V, for example, so as to be ready for regions using an alternating input voltage AC 100 V system, such as, for example, Japan and the United States of America, and regions using an alternating input voltage AC 200 V system, such as, for example, Europe.
As the above-described resonant converter, a resonant converter configured to achieve stabilization by controlling the switching frequency of a switching device forming the converter (a switching frequency control system) is known.
In a resonant converter of such a switching frequency control system configured to switching-drive a switching device using a general-purpose oscillation and drive circuit IC, for example, a maximum variable range of switching frequency fs is, for example, fs=about 50 kHz to about 250 kHz. With such a variable range, under a load condition in which the load power Po is varied in a relatively wide variation range of Po=0 W to about 90 W or even about 150 W, for example, it is substantially impossible to achieve stabilization while dealing with a wide range of alternating input voltage of AC 85 V to 288 V.
When performing stabilization dealing with a variation of load power Po=150 W to 0 W in a single range configuration of an AC 100 V system dealing with variations in a range of about AC 85 V to 144 V, for example, a switching frequency variable range is about 80 kHz to about 200 kHz. To realize a wide range configuration dealing with variations in a range of AC 85 V to 288 V as described above requires a wider range of about 80 kHz to 500 kHz, for example, as a switching frequency control range. With a maximum variable range of the above-mentioned oscillation and drive IC, it is substantially impossible to control the switching frequency.
Even if the oscillation and drive IC can realize such a wide frequency control range, a high switching frequency of 400 kHz or 500 kHz involves an increase in losses at the switching device and the transformer, for example, thus making it very difficult to obtain a practical numerical value as the power conversion efficiency.
For these reasons, it is substantially impossible for resonant converters to be configured to deal with the wide range by switching frequency control.
Accordingly, as a conventional technique enabling realization of a wide range-ready configuration in a resonant converter and power factor improvement, a method using a so-called active filter is known (see Japanese Patent Laid-Open No. Hei 6-327246, for example). The fundamental configuration of such an active filter is shown in FIG. 33 of that reference.
In FIG. 33, a bridge rectifier circuit Di is connected to a commercial alternating-current power supply line AC. An output capacitor Cout is connected in parallel with the positive electrode/negative electrode lines of the bridge rectifier circuit Di. A rectified output of the bridge rectifier circuit Di is supplied to the output capacitor Cout, whereby a direct-current voltage Vout is obtained as the voltage across the output capacitor Cout. This direct-current voltage Vout is supplied as the input voltage to a load 110, such as a DC-to-DC converter, for example, in a succeeding stage.
As shown in the figure, a configuration for power factor improvement includes an inductor L, a fast recovery type diode D, a resistor Ri, a switching device Q, and a multiplier 111.
The inductor L and the diode D are connected in series with each other and inserted between the positive electrode output terminal of the bridge rectifier circuit Di and the positive electrode terminal of the output capacitor Cout.
The resistor Ri is inserted between the negative electrode output terminal (a primary side ground) of the bridge rectifier circuit Di and the negative electrode terminal of the output capacitor Cout.
In this case, a MOS-FET is selected as the switching device Q. As shown in the figure, the switching device Q is inserted between the primary side ground and a point of connection between the inductor L and the diode D.
The multiplier 111 is connected with a current detection line L1 and a waveform input line Lw as a feedforward circuit, and a voltage detection line Lv as a feedback circuit.
The multiplier 111 detects the level of the rectified current flowing through the negative electrode output terminal of the bridge rectifier circuit Di, which level is input from the current detection line L1.
The multiplier 111 also detects the waveform of the rectified voltage at the positive electrode output terminal of the bridge rectifier circuit Di, which waveform is input from the waveform input line Lw. This is equivalent to detection of a waveform of commercial alternating-current power AC (alternating input voltage) as absolute values.
The multiplier 111 also detects the variation difference of the direct-current input voltage on the basis of the direct-current voltage Vout of the output capacitor Cout which voltage is input from the voltage detection line Lv.
The multiplier 111 outputs a drive signal for driving the switching device Q.
The multiplier 111 first multiplies together the level of the rectified current detected from the current detection line L1 and the variation difference of the direct-current input voltage detected from the voltage detection line Lv as described above. Then, the multiplier 111 generates a current command value of the same waveform as the alternating input voltage VAC based on the result of the multiplication and the waveform of the direct-current input voltage detected from the waveform input line Lw.
Further, the multiplier 111 in this case compares the current command value with an actual level of alternating input current (detected on the basis of the input from the current detection line L1), performs PWM control on a PWM signal according to the difference between the current command value and the actual level of the alternating input current, and generates a drive signal based on the PWM signal. The switching device Q is switching-driven by this drive signal. As a result, the alternating input current is controlled so as to have the same waveform as the alternating input voltage, and the power factor approaches one and is thus improved. Also, in this case, since the current command value generated by the multiplier is controlled so as to be varied in amplitude according to the variation difference of the rectified and smoothed voltages, variations in the rectified and smoothed voltages are suppressed.
FIG. 34A shows an input voltage Vin and an input current Iin input to the active filter circuit shown in FIG. 33. The voltage Vin corresponds to the waveform of the voltage as the rectified output of the bridge rectifier circuit Di, while the current Iin corresponds to the waveform of the current as the rectified output of the bridge rectifier circuit Di. The waveform of the current Iin has the same conduction angle as the rectified output voltage (voltage Vin) of the bridge rectifier circuit Di. This indicates that the waveform of the alternating input current flowing from the commercial alternating-current power supply AC to the bridge rectifier circuit Di has the same conduction angle as the current Iin. That is, a power factor close to one is obtained.
FIG. 34B shows the change in energy (power) Pchg input to and output from the output capacitor Cout. The output capacitor Cout stores energy when the input voltage Vin is high, and releases energy when the input voltage Vin is low, thus maintaining a flow of output power.
FIG. 34C shows the waveform of a charge and discharge current Ichg of the above-described output capacitor Cout. As is understood also from the fact that the charge and discharge current Ichg is in phase with the waveform of the input and output energy Pchg of FIG. 34B, the charge and discharge current Ichg flows so as to correspond to an operation of storing/releasing the energy Pchg by the output capacitor Cout.
Unlike the input current Vin, the charge and discharge current Ichg has substantially the same waveform as the second harmonic of the alternating line voltage (commercial alternating-current power AC). A ripple voltage Vd as a second harmonic component occurs in the alternating line voltage as shown in FIG. 34D due to the flow of energy to and from the output capacitor Cout. Because of reactive power conservation, the ripple voltage Vd has a 90° phase difference with respect to the charge and discharge current Ichg shown in FIG. 34C. A rating of the output capacitor Cout is determined in consideration of a second harmonic ripple current and the processing of a high-frequency ripple current from a boost converter switch modulating the second harmonic ripple current.
FIG. 35 shows an example of the configuration of an active filter having a basic control circuit system, which filter is based on the circuit configuration of FIG. 33. Incidentally, the same parts as in FIG. 33 are identified by the same reference numerals, and a description thereof will be omitted.
A switching pre-regulator 115 is disposed between the positive electrode output terminal of a bridge rectifier circuit Di and the positive electrode terminal of an output capacitor Cout. The switching pre-regulator 115 is a part formed by the switching device Q, the inductor L, the diode D and the like in FIG. 33.
The control circuit system including a multiplier 111 also has a voltage error amplifier 112, a divider 113, and a squaring unit 114.
The voltage error amplifier 112 divides a direct-current voltage Vout of the output capacitor Cout by voltage dividing resistors Rvo and Rvd, and then inputs the result to the non-inverting input of an operational amplifier 112a. A reference voltage Vref is input to the inverting input of the operational amplifier 112a. The operational amplifier 112a amplifies a voltage having a level corresponding to an error between the reference voltage Vref and the divided direct-current voltage Vout with an amplification factor determined by a feedback resistor Rv1 and a capacitor Cv1. The operational amplifier 112a then outputs the amplified voltage as an error output voltage Vvea to the divider 113.
The squaring unit 114 is supplied with a so-called feedforward voltage Vff. The feedforward voltage Vff is an output (average input voltage) obtained by averaging an input voltage Vin by an averaging circuit 116 (Rf11, Rf12, Rf13, Cf11, and Cf12). The squaring unit 114 squares the feedforward voltage Vff, and then outputs the squared feedforward voltage Vff to the divider 113.
The divider 113 divides the error output voltage Vvea from the voltage error amplifier 112 by the squared value of the average input voltage which value is output from the squaring unit 114. The divider 113 then outputs a signal resulting from the division to the multiplier 111.
That is, a voltage loop is formed by a system of the squaring unit 114, the divider 113, and the multiplier 111. The error output voltage Vvea output from the voltage error amplifier 112 is divided by the square of the average input voltage (Vff) before being multiplied by a rectified input signal Ivac by the multiplier 111. This circuit maintains the gain of the voltage loop at a constant level without changing the gain as the square of the average input voltage (Vff). The average input voltage (Vff) has an open-loop correcting function, being fed in a forward direction within the voltage loop.
The multiplier 111 is supplied with the output obtained by dividing the error output voltage Vvea by the square of the average input voltage (Vff) and with the rectified output (Iac) of the positive electrode output terminal (a rectified output line) of the bridge rectifier circuit Di via a resistor Rvac. In this case, the rectified output is represented as current (Iac) rather than voltage. The multiplier 111 multiplies these inputs together, thereby generating a current programming signal (multiplier output signal) Imo, and then outputs the current programming signal. This signal corresponds to the current command value described with reference to FIG. 33. The output voltage Vout is controlled by varying the average amplitude of the current programming signal. That is, a PWM signal corresponding to a change in the average amplitude of the current programming signal is generated, and switching driving is performed by a drive signal based on the PWM signal, whereby the level of the output voltage Vout is controlled.
Hence, the current programming signal has a waveform of the average amplitude for controlling the input voltage and the output voltage. Incidentally, the active filter controls not only the output voltage Vout but also the input voltage Vin. It can be said that a current loop in a feedforward circuit is programmed by a rectification line voltage, and therefore the input to a converter (a load 110) in a following stage is resistive.
FIG. 36 shows an example of the configuration of a power supply circuit formed by connecting a current resonant converter in a stage subsequent to an active filter based on the configuration shown in FIG. 33. The power supply circuit shown in this figure is a so-called wide range-ready power supply circuit dealing with alternating input voltages of both an AC 100 V system and an AC 200 V system. Also, the power supply circuit is configured to meet a condition of load power=0 to 150 W. The current resonant converter has the configuration of an externally excited current resonant converter of a half-bridge coupling system.
In the power supply circuit shown in FIG. 36, a common mode noise filter formed by two common mode choke coils CMC and three across capacitors CL are connected to a commercial alternating-current power supply AC in the connection mode shown in the figure, and a bridge rectifier circuit Di is connected in a subsequent stage.
A rectified output line of the bridge rectifier circuit Di is connected with a normal mode noise filter 125 formed by connecting one choke coil LN and two filter capacitors (film capacitors) CN and CN as shown in the figure.
The positive electrode output terminal of the bridge rectifier circuit Di is connected to the positive electrode terminal of a smoothing capacitor Ci via a series connection of the choke coil LN, an inductor LPC of a power choke coil PCC, and a fast recovery type rectifier diode D20. The smoothing capacitor Ci corresponds to the output capacitor Cout in FIG. 33 and FIG. 35. The inductor LPC of the power choke coil PCC and the diode D20 correspond to the inductor L and the diode D, respectively, shown in FIG. 33.
In addition, an RC snubber circuit formed by a capacitor Csn and a resistor Rsn is connected in parallel with the rectifier diode D20 in FIG. 36.
A switching device Q11 in this figure corresponds to the switching device Q10 in FIG. 33. That is, in actually mounting the switching device of the active filter, the switching device Q11 in this case is inserted between a point of connection between the inductor LPC and the fast recovery type rectifier diode D20 and a primary side ground (via a resistor R3). A MOS-FET is selected as the switching device Q11 in this case.
A power factor and output voltage controlling IC 120 in this case is an integrated circuit (IC) that controls the operation of the active filter for improving a power factor so as to approximate the power factor to one.
In this case, the power factor and output voltage controlling IC 120 includes, for example, a multiplier, a divider, a voltage error amplifier, a PWM control circuit, and a drive circuit for outputting a drive signal for switching-driving the switching device. Circuit parts corresponding to the multiplier 111, the voltage error amplifier 112, the divider 113, the squaring unit 114 and the like shown in FIG. 35 are included within the power factor and output voltage controlling IC 120.
In this case, a feedback circuit is formed so as to input a voltage value obtained by dividing the voltage across the smoothing capacitor Ci (rectified and smoothed voltage Ei) by voltage dividing resistors R5 and R6 to a terminal T1 of the power factor and output voltage controlling IC 120.
A feedforward circuit inputs a level of rectified current from a point of connection of a resistor R3 inserted between a source of the switching device Q11 and the primary side ground through a resistor R4 to a terminal T2 of the power factor and output voltage controlling IC 120. That is, a feedforward circuit as a line corresponding to the current detection line L1 in FIG. 33 is formed.
A terminal T4 of the power factor and output voltage controlling IC 120 is supplied with operating power for the power factor and output voltage controlling IC 120. A half-wave rectifier circuit formed by a diode D21 and a capacitor 21 shown in the figure converts an alternating voltage excited in a winding N5 coupled to the inductor LPC by transformer coupling in the power choke coil PCC into a low direct-current voltage, and then supplies the low direct-current voltage to the terminal T4 of the power factor and output voltage controlling IC 120.
The power factor and output voltage controlling IC 120 outputs, from a terminal T3 thereof, a drive signal for driving the switching device to a gate of the switching device Q11.
The switching device Q11 performs a switching operation according to the drive signal applied thereto.
As described with reference to FIG. 33 and FIG. 35, the switching device Q11 is switching-driven by the drive signal based on PWM control such that the conduction angle of the rectified output current is substantially equal to the conduction angle of the waveform of the rectified output voltage. The conduction angle of the rectified output current being substantially equal to the conduction angle of the waveform of the rectified output voltage means that the conduction angle of an alternating input current flowing in from the commercial alternating-current power supply AC is substantially equal to the conduction angle of the waveform of an alternating input voltage VAC. Consequently, the power factor is controlled to be substantially one. That is, the power factor is improved.
As figures representing the actual power factor improving operation, FIG. 37 and FIG. 38 show waveforms of the alternating input current IAC obtained in the circuit shown in FIG. 36 together with the alternating input voltage VAC by way of comparison. FIG. 37 shows a result when the alternating input voltage VAC=100 V. FIG. 38 shows a result when the alternating input voltage VAC=230 V.
As shown in FIG. 37, the peak level of the alternating input current IAC is 6.5 Ap when the alternating input voltage VAC=100 V. It is understood that the conduction period of the alternating input current IAC substantially coincides with the conduction period of the alternating input voltage VAC and that the power factor is thereby improved.
When the alternating input voltage VAC=230 V, as shown in FIG. 38, the peak level of the alternating input current IAC is 3.0 Ap. It is to be understood that, also in this case, the conduction period of the alternating input current IAC substantially coincides with the conduction period of the alternating input voltage VAC and that the power factor is thereby improved.
In addition to such power factor improvement, the power factor and output voltage controlling IC 120 shown in FIG. 36 operates so as to make constant the average value of the rectified and smoothed voltage Ei (corresponding to Vout in FIG. 35)=380 V in the range of the alternating input voltage VAC=85 V to 264 V. That is, the direct-current input voltage stabilized at 380 V is supplied to the current resonant converter in the subsequent stage regardless of the variation range of the alternating input voltage VAC=85 V to 264 V. This is also indicated by a decrease in the peak level of the alternating input current IAC to less than ½ when the alternating input voltage VAC=230 V in FIG. 37 and FIG. 38.
Such a range of the alternating input voltage VAC=85 V to 264 V continuously covers the commercial alternating-current power supply AC 100 V system and the commercial alternating-current power supply AC 200 V system. Hence, the switching converter in the subsequent stage is supplied with the direct-current input voltage (Ei) stabilized at the same level in the cases of the commercial alternating-current power supply AC 100 V system and the commercial alternating-current power supply AC 200 V system. That is, having the active filter, the power supply circuit shown in FIG. 36 is also configured as a wide range power supply circuit.
The current resonant converter in the stage subsequent to the active filter includes two switching devices Q1 and Q2 as shown in the figure. In this case, the switching devices Q1 and Q2 are connected to each other by half-bridge coupling such that the switching device Q1 is on a high side and the switching device Q2 is on a low side. The switching devices Q1 and Q2 are connected in parallel with the rectified and smoothed voltage Ei (direct-current input voltage). That is, the converter forms a current resonant converter of a half-bridge coupling system.
The current resonant converter in this case is externally excited. Correspondingly, a MOS-FET is used as the switching devices Q1 and Q2. Clamping diodes DD1 and DD2 are connected in parallel with the switching devices Q1 and Q2, respectively, whereby a switching circuit is formed. The clamping diodes DD1 and DD2 form a path for passing a current in an opposite direction when the switching devices Q1 and Q2 are turned off.
The switching devices Q1 and Q2 are switching-driven by an oscillation and drive circuit 2 at a required switching frequency in timing in which the switching devices Q1 and Q2 are turned on/off alternately. The oscillation and drive circuit 2 operates so as to variably control the switching frequency under control according to the level of a secondary side direct-current output voltage Eo to be described later by a control circuit 1 shown in the figure. The oscillation and drive circuit 2 thereby stabilizes the secondary side direct-current output voltage Eo.
An isolated converter transformer PIT is provided to transmit the switching output of the switching devices Q1 and Q2 from the primary side to the secondary side.
One end part of a primary winding N1 of the isolated converter transformer PIT is connected to a point of connection (a switching output point) between the switching devices Q1 and Q2. Another end part of the primary winding N1 of the isolated converter transformer PIT is connected to the primary side ground via a series resonant capacitor C1. The capacitance of the series resonant capacitor C1 and the leakage inductance (L1) of the primary winding N1 form a series resonant circuit. The series resonant circuit performs a resonant operation by being supplied with the switching output of the switching devices Q1 and Q2. The series resonant circuit thereby converts the operation of the switching circuit formed by the switching devices Q1 and Q2 into a current resonance type operation.
A secondary winding N2 is wound on the secondary side of the isolated converter transformer PIT. The secondary winding N2 in this case is connected with a full-wave rectifier circuit formed by a bridge rectifier circuit including rectifier diodes Do1 to Do4 connected to each other by a bridge connection as shown in the figure and a smoothing capacitor Co. A secondary side direct-current output voltage Eo is thereby obtained as the voltage across the smoothing capacitor Co. This secondary side direct-current output voltage Eo is supplied to a load side not shown in the figure, and also branches off to be input as a detection voltage for the above-described control circuit 1. The control circuit 1 supplies a control signal corresponding to the level of the secondary side direct-current output voltage Eo input to the control circuit 1 to the oscillation and drive circuit 2. The oscillation and drive circuit 2 drives the switching devices Q1 and Q2 such that the switching frequency of the switching devices Q1 and Q2 is varied to stabilize the secondary side direct-current output voltage Eo according to the control signal. That is, the secondary side direct-current output voltage Eo is stabilized by a switching frequency control system.
FIG. 39 shows the characteristics of AC→DC power conversion efficiency (total efficiency), the power factor, and the rectified and smoothed voltage Ei with respect to the load variation. This figure shows the characteristics with respect to the variation of load power Po=150 W to 0 W. The characteristics at the time of the alternating input voltage VAC=100 V (AC 100 V system) are represented by solid lines, and the characteristics at the time of the alternating input voltage VAC=230 V (AC 200 V system) are represented by broken lines.
FIG. 40 shows the characteristics of AC→DC power conversion efficiency (total efficiency), the power factor, and the rectified and smoothed voltage Ei with respect to the variation of the alternating input voltage VAC. This figure shows the characteristics with respect to the variation of the alternating input voltage VAC=85 V to 264 V under a fixed condition of load power Po=150 W.
As shown in FIG. 39, the AC→DC power conversion efficiency (ηAC→DC) is increased as the load power Po is increased. With respect to the variation in the alternating input voltage VAC, under the same load condition, the AC→DC power conversion efficiency (ηAC→DC) is increased as the level of the alternating input voltage VAC is raised, as shown in FIG. 39 and FIG. 40.
In practice, under the load condition of the load power Po=150 W, ηAC→DC=about 88.0% when the alternating input voltage VAC=100 V, and ηAC→DC=about 91.0% when the alternating input voltage VAC=230 V.
As shown in FIG. 39, the power factor PF is increased as the load power Po is increased. As shown in FIG. 39 and FIG. 40, with respect to the variation of the alternating input voltage VAC, the power factor PF is decreased as the level of the alternating input voltage VAC is raised.
In practice, under the load condition of the load power Po=150 W, the power factor PF=about 0.99 when the alternating input voltage VAC=100 V, and the power factor PF=about 0.98 when the alternating input voltage VAC=230 V.
As shown in FIG. 39 and FIG. 40, the rectified and smoothed voltage Ei is constant against variations of the load power Po=150 W to 0 W and the alternating input voltage VAC=85 V to 264 V.
As is understood from the description thus far, the power supply circuit shown in FIG. 36 is formed with the conventionally known active filter shown in FIG. 33 and FIG. 35. The power factor is improved by employing such a configuration. In addition, under the condition of a load power of 150 W or lower, a so-called wide range-ready configuration operating in the commercial alternating-current power supply AC 100 V system and the commercial alternating-current power supply AC 200 V system is realized.
However, the power supply circuit shown in FIG. 36 has the following problems.
First, the power conversion efficiency of the power supply circuit shown in FIG. 36 is a total of the AC→DC power conversion efficiency corresponding to the active filter in the preceding stage and the DC→DC power conversion efficiency of the current resonant converter in the succeeding stage, as is also shown in the figure.
That is, the total power conversion efficiency of the circuit shown in FIG. 36 is a value obtained by multiplying the values of these power conversion efficiencies together, and thus tends to be correspondingly decreased.
According to an experiment, the AC→DC power conversion efficiency of a part corresponding to the active filter in the circuit of FIG. 36 is ηAC→DC=about 93% when the alternating input voltage VAC=100 V, and is ηAC→DC=about 96% when the alternating input voltage VAC=230 V. The DC→DC power conversion efficiency on the current resonant converter side is ηDC→DC=about 95% when the load power Po=150 W and the rectified and smoothed voltage Ei=380 V.
Thus, the total AC-to-DC power conversion efficiency of the circuit of FIG. 36 is decreased to ηAC→DC=about 88.0% when the alternating input voltage VAC=100 V, and is decreased to q AC→DC=about 91.0% when the alternating input voltage VAC=230 V, as described with reference to FIG. 39 and FIG. 40.
In addition, since the active filter circuit performs a hard switching operation, a very high level of noise occurs. Therefore, a relatively serious measure to suppress noise is required.
Thus, in the circuit shown in FIG. 36, the noise filter formed by the two common mode noise choke coils and the three across capacitors is formed in the line of the commercial alternating-current power supply AC. That is, a filter of two or more stages is required.
Also, the normal mode noise filter formed by the one choke coil LN and the two filter capacitors CN is provided in the rectified output line. Further, the RC snubber circuit is provided for the fast recovery type rectifier diode D20 for rectification.
Thus, the actual circuit requires measures against noise using a very large number of parts, which results in an increase in cost and an increase in the mounting area of the power supply circuit board.
Furthermore, while the switching frequency of the switching device Q11 operated by the power factor and output voltage controlling IC 120 as a general-purpose IC is fixed at 60 kHz, the switching frequency of the current resonant converter in the subsequent stage is varied in a range of 80 kHz to 200 kHz. Since switching timings of the switching device Q11 and the current resonant converter are thus independent of each other, primary side ground potentials obtained by switching operations of the switching device Q11 and the current resonant converter interfere with each other and thus become unstable, so that an abnormal oscillation, for example, tends to occur. This, for example, makes circuit design difficult and degrades reliability.